Apparatus and method for substantially eliminating a near-channel interfering amplitude modulated signal

ABSTRACT

An apparatus and method are provided that compensates for the problematic time-varying DC offset by effectively eliminating a near-channel amplited modulated interferer from a signal. The apparatus includes a first channel estimator for estimating a plurality of first channel filter taps Ĥ using a first signal model S t , and a second channel estimator for estimating a plurality of second channel filter taps {tilde over (H)} using a second signal model {tilde over (S)} t . The apparatus also includes a processor for selecting which of the first signal model S t  and the second signal model {tilde over (S)} t  is to be used or was used to substantially eliminate the near-channel amplitude modulated interferer from the received signal. The apparatus can be a mobile phone, base station, direct conversion receiver, or communications system (for example).

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of prior U.S. patent application Ser.No. 09/398,668, filed Sep. 17, 1999, incorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Technical Field of the Invention

The present invention generally relates to the telecommunications fieldand, in particular, to an apparatus and method that compensates for aproblematic time-varying DC offset by effectively eliminating anear-channel interfering amplitude modulated (AM) signal from acommunications channel.

2. Description of Related Art

In the telecommunications field, one of the most significant designchallenges involves the development of new direct conversion receiversthat are capable of improving the demodulated quality of a signal.Traditional direct conversion receivers or homodyne receivers generallyoperate to demodulate an incoming signal by directly converting theincoming signal down to baseband, without the use of any intermediatefrequencies, and outputting a desired signal. An example of thetraditional direct conversion receiver is briefly discussed below withrespect to FIG. 1.

Referring to FIG. 1 (PRIOR ART), there is illustrated a block diagram ofa traditional direct conversion receiver 100. Basically, the traditionaldirect conversion receiver 100 includes an antenna 102 for receiving asignal from a transmitter 104. The received signal is filtered by a bandpass filter (BPF) 106 designed to pass a desired frequency band such asthe GSM (Global System for Mobile Communications) frequency band fromthe received signal. The filtered signal is amplified in a low noiseamplifier (LNA) 108 and down-converted to a base band Inphase (I)component and a base band Quadrature (Q) component using mixers 114 aand 114 b, respectively, and a local oscillator (LO) 116. The localoscillator 116 outputs a frequency adapted to a carrier frequency of thereceived signal. The base band I and Q components are respectivelyfiltered by first low pass filters (LPFs) 118 a and 118 b, converted todigital signals by analog-to-digital convertors (A/Ds) 120 a and 120 b,and then filtered by second low pass filters (LPFs) 122 a and 122 b toobtain a signal format that can be handled by a data recovery unit (DR)124. The data recovery unit 124 operates to demodulate the receivedsignal.

Traditional direct conversion receivers 100 have an efficient radioreceiver architecture in terms of cost, size and current consumption.However, traditional direct conversion receivers 100 suffer from thewell known DC offset problem that can be attributable to three differentsources: (1) transistor mismatch in a signal path; (2) the localoscillator 116 outputting a signal that leaks and self-down converts toDC when passed through mixers 114 a and 114 b; and (3) a largenear-channel amplitude modulated (AM) interfering signal leaking intothe local oscillator 116 and self-downconverting to DC. Since, theresulting DC offset can be several decibels (dB) larger than theinformation signal, one should take care of the DC offset to be able torecover the transmitted data in the data recovery unit 124.

The DC offsets due to (1) and (2) can be assumed to be constant duringone burst (i.e., a number of received symbols) and can be taken care ofby adding an extra DC component to the signal model used whiledemodulating the transmitted data in the data recovery unit 124. Thismethod is well known in the art. However, the DC offset due to (3) istime-varying because of the amplitude variations in the interferingsignal and as such it is difficult to compensate for this particular DCoffset. Two examples of how the traditional direct conversion receiver100 can be adapted to compensate for such AM interfering signals aredisclosed in WO 98/04050 and EP 0 806 841, and briefly described belowwith respect to FIG. 2.

Referring to FIG. 2 (PRIOR ART), there is illustrated a block diagram ofa traditional direct conversion receiver 200 configured to compensatefor AM interfering signals as described in WO 98/04050 and EP 0 806 841.The general idea disclosed in both of these documents is to add a thirdreceiver 202 (in addition to the I and Q receivers described above)designed to compensate for the dominating AM interfering signal.

The traditional direct conversion receiver 200 excluding the thirdreceiver 202 generally operates as the direct conversion receiver 100described above wherein like numerals represent like parts throughoutFIGS. 1 and 2. For purposes of the discussion related to the directconversion receiver 200 of FIG. 2, the received signal can include awanted signal y_(t) and an unwanted near-channel interferer p_(t). Dueto nonlinear effects in the low noise amplifier 108 and the mixer 114 ait can be shown that the dominated output from the second low-passfilter 122 a is a wanted I component I_(t) and a fraction of the squaredenvelope of the interfering signal a|p_(t)|². Likewise, the dominatedoutput from the second low-pass filter 122 b is a wanted Q componentQ_(t) and a fraction of the squared envelope of the interfering signalb|p_(t)|².

The third receiver 202 is designed to take into account the nonlineareffects within the low noise amplifier 108 and the mixers 114 a and 114b which collectively operate to convert the interfering signal to a baseband signal. The low noise amplifier 108 directs the received signal toa power detector (PD) 204 which functions to detect an envelope of thereceived signal. It should be noted that this detected envelope consistsmainly of the envelope attributable to the near-channel AM interferingsignal whenever the unwanted interferer p_(t) is much larger than thewanted signal y_(t). The power detected signal is then converted intothe digital domain by an analog-to-digital convertor (A/D) 206, filteredby a low pass filter (LPF) 208 and fed to a control unit (CU) 210 whichmultiplies the detected envelopes with estimated parameters â and{circumflex over (b)}. The estimated interfering signals â|p_(t)|² and{circumflex over (b)}|p_(t)|² of the distortion are respectively inputto subtractors 212 a and 212 b and subtracted from the I and Qcomponents to obtain “relatively clean” I and Q components. The“relatively clean” I and Q components are then input to the datarecovery unit 124.

Even if the solution to the DC offset problem described in WO 98/04050and EP 0 806 841 appears to be promising it still has disadvantages, interms of cost and current, due to the need to implement a thirdreceiver. Therefore, there is a need for an apparatus and method thatcan suppress the near-channel AM interferer in a cost and currentefficient manner.

BRIEF DESCRIPTION OF THE INVENTION

The present invention is an apparatus and method that compensates forthe problematic time-varying DC offset by substantially eliminating anear-channel amplitude modulated interferer from a signal. The apparatusincludes a first channel estimator for estimating a plurality of firstchannel filter taps Ĥ using a first signal model S_(t), and a secondchannel estimator for estimating a plurality of second channel filtertaps {tilde over (H)} using a second signal model {tilde over (S)}_(t).The apparatus also includes a processor for selecting which of the firstsignal model S_(t) and the second signal model {tilde over (S)}_(t) isto be used or was used to substantially eliminate the near-channelamplitude modulated interferer from the received signal. The apparatusand method can be implemented in, for example, a mobile phone, basestation, direct conversion receiver, or communications system.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the method and apparatus of the presentinvention may be had by reference to the following detailed descriptionwhen taken in conjunction with the accompanying drawings wherein:

FIG. 1 (PRIOR ART) is a block diagram illustrating the basic componentsof a traditional direct conversion receiver;

FIG. 2 (PRIOR ART) is a block diagram illustrating a traditional directconversion receiver adapted to include a third receiver for DC offsetcompensation;

FIG. 3 is a block diagram illustrating the basic components associatedwith an exemplary apparatus of the present invention;

FIG. 4 is a block diagram illustrating in greater detail a firstembodiment of the exemplary apparatus shown in FIG. 3;

FIG. 5 is a block diagram illustrating in greater detail a secondembodiment of the exemplary apparatus shown in FIG. 3;

FIG. 6 is a block diagram illustrating in greater detail a thirdembodiment of the exemplary apparatus shown in FIG. 3;

FIG. 7 is a block diagram illustrating in greater detail a fourthembodiment of the exemplary apparatus shown in FIG. 3; and

FIG. 8 is a flowchart illustrating the basic steps of the preferredmethod in accordance with the present invention.

DETAILED DESCRIPTION OF THE DRAWINGS

Referring to FIGS. 3-8, there are disclosed exemplary embodiments of anapparatus 300 and preferred method 800 each of which is capable ofeffectively eliminating near-channel amplitude modulated (AM)interfering signals from signals in accordance with the presentinvention.

Although the apparatus 300 is described with respect to a directconversion receiver (e.g., homodyne receiver) used in a digitalcommunications system, it should be understood that the presentinvention can be used within any communications device, but isespecially suited for use with a mobile phone and base station.Accordingly, the different embodiments of the apparatus 300 andpreferred method 800 should not be construed in a limited manner.

Referring to FIG. 3, there is a block diagram illustrating the basiccomponents associated with the exemplary apparatus 300 of the presentinvention. Basically, the apparatus 300 (described as a directconversion receiver) enables the suppression of AM interfering signalsusing only the I and Q baseband signals in a predetermined manner suchthat no extra receiver (see the third receiver 202 of FIG. 2) is needed,implying a cost and current efficient receiver having low complexity andhigh performance.

More specifically, the direct conversion receiver 300 of the presentinvention includes a first channel estimator 302 operable to estimate aplurality of first channel filter taps Ĥ using a first signal modelS_(t), and a second channel estimator 304 operable to estimate aplurality of second channel filter taps {tilde over (H)} using a secondsignal model {tilde over (S)}_(t). The direct conversion receiver 300also includes a processor 306 operable to select either the first signalmodel S_(t) or the second signal model {tilde over (S)}_(t) that is tobe used or was used to substantially eliminate the near-channel AMinterfering signal from the received signal. A detailed description ofhow each embodiment of the direct conversion receiver 300 operates tosubstantially eliminate the AM interfering signal is provided below withrespect to FIGS. 4-7.

Referring to FIG. 4, there is a block diagram illustrating in greaterdetail a first embodiment of an exemplary direct conversion receiver 400in accordance with the present invention. Certain details associatedwith direct conversion receivers are known in the industry and as suchneed not be described herein. Therefore, for clarity, the descriptionsprovided below in relation to the direct conversion receivers of thepresent invention omit some elements known to those skilled in the artthat are not necessary to understand the invention.

The direct conversion receiver 400 includes an antenna 402 for receivinga signal from a transmitter 404. The received signal is filtered by aband pass filter (BPF) 406 designed to pass a desired frequency bandsuch as the GSM (Global System for Mobile Communications) frequency bandfrom the received signal. The received signal output from the band passfilter 406 can be represented as: $\begin{matrix}{w_{t} = {y_{t} + {\overset{\sim}{p}}_{t}}} & (1) \\{\quad{= {{r_{t}\quad\cos\quad\left( {{\omega_{o}t} + \varphi_{t}} \right)} + {p_{t}\quad\cos\quad\left( {{\omega_{1}t} + \theta_{t}} \right)}}}} & (2)\end{matrix}$where w_(t) is the received signal, y_(t) is the desired signal atcarrier frequency ω₀, {tilde over (p)}_(t) is the near-channel AMinterfering signal at carrier frequency ω₁, and ω₀ and ω₁ are within thepass band of the band pass filter 406.

The filtered signal is amplified in a low noise amplifier (LNA) 408 anddown-converted to a base band Inphase (I) component and a base bandQuadrature (Q) component using mixers 414 a and 414 b, respectively, anda local oscillator (LO) 416. The local oscillator 416 outputs twooscillation signals LO_(I) and LO_(Q) adapted to a carrier frequency ofthe received signal, the two oscillation signals can be represented asfollows:LO _(I)(t)=cos(ω₀ t)  (3)LO _(Q)(t)=sin(ω₀ t)  (4)where LO_(I) and LO_(Q) are the oscillating signals associated with theI and Q components, respectively. The oscillating signals LO_(I) andLO_(Q) and the received signal are multiplied in the mixers 414 a and414 b.

Due to the nonlinearities of the local oscillator 416 and interfererleakage (represented with the scale factors α′, β′), the low passfiltered signal output from the I component mixer 414 a and a first lowpass filter (LPF) 418 a can be represented as follows: $\begin{matrix}{{\overset{\sim}{I}}_{t} = {{LPF}\left\{ {\left( {{r_{t}\cos\left( {{\omega_{0}t} + \varphi_{t}} \right)} + {p_{t}{\cos\left( {{\omega_{1}t} + \theta_{t}} \right)}} + {\alpha_{I}^{'}{\cos\left( {{\omega_{0}t} + \gamma} \right)}}} \right)*\quad(5)} \right.}} \\\left. \left( {{\cos\left( {\omega_{0}t} \right)} + {\beta_{I}^{'}p_{t}{\cos\left( {{\omega_{1}t} + \theta_{t} + \delta} \right)}}} \right) \right\} \\{= {{r_{t}\quad\cos\quad\varphi_{t}} + {\beta_{I}p_{t}^{2}} + {\alpha_{I}\quad(6)}}}\end{matrix}$Likewise, the low pass filtered signal output from the Q component mixer414 b and a first low-pass filter (LPF) 418 b can be represented asfollows:{tilde over (Q)} _(t) =r _(t) sin φ_(t)+β_(Q) p _(t) ²+α_(Q)  (7)

Thereafter, the Ĩ and {tilde over (Q)} components are respectivelyconverted into digital domain by analog-to-digital convertors (A/Ds) 420a and 420 b and respectively filtered by second low pass filters (LPFs)422 a and 422 b. And, after certain normalizations the base bandcomponents can be represented as:Ĩ _(t) =I _(t) +a|p _(t)|² +I _(DC)  (8){tilde over (Q)} _(t) =Q _(t) +b|p _(t)|² +Q _(DC)  (9)where I_(t), Q_(t) are the wanted I and Q components, and I_(DC), Q_(DC)are the DC components on the I and Q components, respectively. |p_(t)|²is the low pass filtered and sampled squared envelope of the interferingAM signal. In case of digital transmission over radio channels withintersymbol interference, such as for instance in GSM or a D-AMPScellular systems, the wanted I and Q components can be written in acomplex notation as follows:I _(t) +jQ _(t) =H ^(T) U _(t) +e _(t)  (10)where H=[h₀, . . . , h_(L)]^(T) is a vector of complex valued channelfilter taps, U_(t)=[u_(t), . . . , u_(t-L)]^(T) is a vector of complextransmitted symbols, and e_(t) is some kind of complex valued noise.Therefore, the complex valued base band signal for a first signal modelS_(t) can be represented as: $\begin{matrix}{S_{t} = {{\overset{\sim}{I}}_{t} + {j{\overset{\sim}{Q}}_{t}}}} & (11) \\{\quad{= {{H^{T}U_{t}} + {\left( {a + {j\quad b}} \right){p_{t}}^{2}} + {DC} + e_{t}}}} & (12) \\{\quad{= {{\sum\limits_{k = o}^{L}\quad{\left\{ {h_{k}^{I} + {j\quad h_{u}^{Q}}} \right\}\left\{ {U_{t}^{I} + {j\quad U_{t}^{Q}}} \right\}}} + e_{t}^{I} + {j\quad e_{t}^{Q}}}}} & (13)\end{matrix}$where j=√{square root over (−1)}.

The Ĩ_(t) and {tilde over (Q)}_(t) components are input to a firstchannel estimator 424 a that correlates, using the first signal modelS_(t), a known training sequence (TS) with the received signal S_(t)(which contains the same known training sequence) to determine asynchronization position and an estimate of the first set of channelfilter taps Ĥ. The use of the first signal model S_(t) in estimating thechannel filter taps Ĥ is well known in the art. In fact, the firstsignal model S_(t) was the only signal model used to estimate channelfilter taps Ĥ in traditional direct conversion receivers. The estimatedchannel filter taps Ĥ are input to a processor 426 that is described indetail below.

Generally, the present invention includes a second channel estimator 424b that uses a second signal model {tilde over (S)}_(t) to estimate asecond set of channel filter taps {tilde over (H)} that are input to theprocessor 426 which selects the signal model S_(t) or {tilde over(S)}_(t) that is to be used to further process the received signal. Inother words, the processor 426 selects the appropriate signal modelS_(t) or {tilde over (S)}_(t) using the estimated channel filter taps Ĥand {tilde over (H)} and some other parameters (e.g., residuals)discussed below. Thereafter, an equalizer 428, coupled to the processor426, uses either the first or second set of channel filter taps Ĥ or{tilde over (H)} corresponding to the selected signal model S_(t) or{tilde over (S)}_(t) to equalize the received signal. It should be notedthat the equalizer 428 also receives It from the second low pass filter422 a, {tilde over (Q)}_(t) from the second low pass filter 422 b, and${\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}$from a subtractor 434 before equalizing the received signal.

More specifically, the second signal model {tilde over (S)}_(t),represented as the component${\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}$where â and {circumflex over (b)} are respective estimates of a and bfrom equations 8 and 9, is input to the second channel estimator 424 b.The second channel estimator 424 b operates to correlate the knowntraining sequence (TS) with the${\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}$component to determine a synchronization position and an estimate of thesecond set of channel filter taps {tilde over (H)}. The$\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}$portion of the second signal model {tilde over (S)}_(t) is generatedusing a multiplicator 432 which receives Ĩ_(t) from the second low passfilter 422 a, and receives the estimated parameters â and {circumflexover (b)} from the processor 426. The multiplicator 432 outputs$\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}$to the subtractor 434 which receives {tilde over (Q)}_(t) from thesecond low pass filter 422 b and outputs the component${\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{{\overset{\sim}{I}}_{t}.}}$

The second signal model {tilde over (S)}_(t) can be represented asfollows: $\begin{matrix}{{{\overset{\sim}{S}}_{t}\left( {\hat{b}/\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}\quad{\overset{\sim}{I}}_{t}}}} & (14) \\{\quad{= {{{\overset{\sim}{H}}^{T}{\overset{\sim}{U}}_{t}} + {{imag}\quad\left( e_{t} \right)} - {\frac{\hat{b}}{\hat{a}}\quad{real}\quad\left( e_{t} \right)} + R_{D\quad C}}}} & (15)\end{matrix}$where Ũ_(t)=[real(U_(t)) imag(U_(t))] does not contain any distortion|p_(t)|², implying the elimination of the AM interferer.

The derivation to obtain equation (15) from equation (14) follows:$\begin{matrix}{{{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}\quad{\overset{\sim}{I}}_{t}}}} & (16) \\{\quad{= {Q_{t} - {\frac{\hat{b}}{\hat{a}}\quad I_{t}} + {\left\lbrack {b - \frac{\hat{b}a}{\hat{a}}} \right\rbrack{p_{t}}^{2}} + \left\{ {Q_{D\quad C} - {\frac{\hat{b}}{\hat{a}}I_{D\quad C}}} \right\}}}} & \quad\end{matrix}$where${Q_{D\quad C} - {\frac{\hat{b}}{\hat{a}}I_{D\quad C}}} = R_{D\quad C}$(see equation 15), and assuming â=a and {circumflex over (b)}=b then|p_(t)|² vanishes.Equation 10 can be written as: $\begin{matrix}{{I_{t} + {j\quad Q_{t}}} = {{\sum\limits_{k}^{\quad}\quad\left\{ {{h_{k}^{I}u_{t - k}^{I}} - {h_{k}^{Q}u_{t - k}^{Q}}} \right\}} + {j\left\{ {{h_{k}^{I}u_{t - k}^{Q}} + {h_{k}^{Q}u_{t - k}^{I}}} \right\}} + e_{t}^{I} + {j\quad e_{t}^{Q}}}} & (17)\end{matrix}$where e_(t) is expressed in real and imaginary parts as e_(t)=e_(t)^(I)+je_(t) ^(Q), h^(I) and h^(Q) respectively represent the real andimaginary parts of the channel filter taps h, and u^(I) and u^(Q)respectively represent the real and imaginary parts of the transmittedsymbols u_(t).Rearranging equation 17 yields $\begin{matrix}{{I_{t} + {j\quad Q_{t}}} = {e_{t}^{I} + \left\{ {{\sum\limits_{k}{h_{k}^{I}\quad u_{t - k}^{I}}} - {h_{k}^{Q}\quad u_{t - k}^{Q}}} \right\} +}} & (18) \\{\quad{j\left( {\left\{ {{\sum\limits_{k}{h_{k}^{I}\quad u_{t - k}^{Q}}} + {h_{k}^{Q}\quad u_{t - k}^{I}}} \right\} + e_{\quad t}^{Q}} \right)}\quad} & \quad \\{{{{where}\quad e_{t}^{I}} + \left\{ {{\sum\limits_{k}{h_{k}^{I}\quad u_{t - k}^{I}}} - {h_{k}^{Q}\quad u_{t - k}^{Q}}} \right\}} = I_{t}} & \quad \\{{{and}\quad\left( {\left\{ {{\sum\limits_{k}{h_{k}^{I}\quad u_{t - k}^{Q}}} + {h_{k}^{Q}\quad u_{t - k}^{I}}} \right\} + e_{\quad t}^{Q}} \right)}\quad = {\overset{\sim}{Q}}_{t}} & \quad\end{matrix}$Then, substituting for I_(t) and Q_(t) in equation 16, and assuming â=aand {circumflex over (b)}=b, $\begin{matrix}{{{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\sum\limits_{k}{\left( {h_{k}^{Q} - {\frac{\hat{b}}{\hat{a}}h_{k}^{I}}} \right)u_{t - k}^{I}}} + {\sum\limits_{k}{\left( {h_{k}^{I} - {\frac{\hat{b}}{\hat{a}}h_{k}^{Q}}} \right)u_{t - k}^{Q}}} +}} & (19) \\{\quad{{{imag}\quad\left\{ e_{t} \right\}} - {\frac{\hat{b}}{\hat{a}}\quad{re}\quad\left\{ e_{t} \right\}} + R_{D\quad C}}} & \quad\end{matrix}$where imag{e_(t)}=e_(t) ^(I) and re(e_(t))=e_(t) ^(Q).And, in matrix form, $\begin{matrix}{{\overset{\sim}{S}}_{t} = \left\{ {{\left\lbrack {h_{0}^{Q} - {\frac{\hat{b}}{\hat{a}}h_{0}^{I}}} \right\rbrack\quad{\cdots\quad\left\lbrack {h_{L}^{Q} - {\frac{\hat{b}}{\hat{a}}h_{L}^{I}}} \right\rbrack}},{\left\lbrack {h_{0}^{I} - {\frac{\hat{b}}{\hat{a}}h_{0}^{Q}}} \right\rbrack\quad{\cdots\quad\left\lbrack {h_{L}^{I} - {\frac{\hat{b}}{\hat{a}}h_{L}^{Q}}} \right\rbrack}}} \right\}} & (20) \\{\quad{\begin{Bmatrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}u_{t}^{I} \\\vdots\end{matrix} \\u_{t - L}^{I}\end{matrix} \\u_{t}^{Q}\end{matrix} \\\vdots\end{matrix} \\u_{t - L}^{Q}\end{Bmatrix} + {{imag}\quad\left\{ e_{t} \right\}} - {\frac{\hat{b}}{\hat{a}}\quad{re}\quad\left\{ e_{t} \right\}} + R_{D\quad C}}} & \quad\end{matrix}$Equation 20 can now be written in the form of equation 15:$\begin{matrix}{= {{{\overset{\sim}{H}}^{T}{\overset{\sim}{U}}_{t}} + {{imag}\quad\left\{ e_{t} \right\}} - {\frac{\hat{b}}{\hat{a}}\quad{real}\quad\left\{ e_{t} \right\}} + R_{D\quad C}}} & (21)\end{matrix}$

However, since a and b (see equations 8 and 9) are not known, they haveto be estimated within the channel estimator 424 b using the secondsignal model {tilde over (S)}_(t) and the resulting DC component,R_(DC). One way of estimating â and {circumflex over (b)} for eachreceived burst is described in the exemplary optimizing b/a algorithmwhich follows:

-   1. Set i=o.-   2. $\begin{matrix}    {{{Let}\quad\frac{\hat{b}}{\hat{a}}} = {\frac{b_{i}}{a_{i}}.}} & (22)    \end{matrix}$    The start value b₀/a₀ can be based on some a priori information    about these parameters, for instance some nominal nonlinear    performance for the particular low noise amplifier 408 and mixers    414 a and 414 b. For instance, the b/a ratio can be between 1/10 to    10 then one can make a grid of N values (corresponding to i=0 . . .    N−1) between 1/10 to 10 and compute Q (b_(i)/a_(i)) for all of these    values (see Equation 24 and Steps 6-7 below). $\begin{matrix}    {{{Find}\quad{{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)}} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}\quad{\overset{\sim}{I}}_{t}}}} & (23)    \end{matrix}$-   4. Synchronize to find the best synchronization position or known    symbol pattern in the received burst. For example, this can be done    by correlating between the received burst and the training sequence.-   5. Channel estimate to find the estimated channel filter taps {tilde    over (H)} and R_(DC) for the signal    ${{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)}.$    For example, this can be done using conventional least-squares    techniques as indicated by equation 24 below: $\begin{matrix}    {{Q^{\min}\left\lbrack \frac{\hat{b}}{\hat{a}} \right\rbrack} = {\begin{matrix}    \begin{matrix}    \min \\    {{\overset{\sim}{h}}_{k}^{I},{\overset{\sim}{h}}_{k}^{Q},}    \end{matrix} \\    R_{D\quad C}    \end{matrix}{\sum\limits_{L = 1}^{N}\quad\left\lbrack {{{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} - {\sum\limits_{k = 1}^{L}\quad{{\overset{\sim}{h}}_{k}^{I}\quad u_{t - k}^{I}}} - {\sum\limits_{k = 1}^{L}\quad{{\overset{\sim}{h}}_{k}^{Q}u_{t - k}^{Q}}} - R_{D\quad C}} \right\rbrack^{2}}}} & (24) \\    {{{store}\quad{Q^{\min}\left\lbrack \frac{\hat{b}}{\hat{a}} \right\rbrack}} = {{{f(i)}\quad{and}\quad{\overset{\sim}{H}(i)}} = {{\overset{\sim}{H}}_{opt}\left\{ \frac{b_{i}}{a_{i}} \right\}}}} & \quad    \end{matrix}$    where {tilde over (H)}_(opt) is the vector that yields f(i) in    equation 24.-   6. Set i=i+1-   7. Perform steps 2-6 repetitively until all $\frac{a_{i}}{b_{i}}$    (e.g., i=0 . . . N−1) are used.-   8. Find the lowest value, f_(min), of all of the f(i) values. Select    the corresponding $\frac{a_{i}}{b_{i}}$    value to be $\frac{\hat{a}}{\hat{b}},$    and select the corresponding {tilde over (H)}(i) vector to be the    estimated channel tap vector {tilde over (H)}.

Another way of estimating â and {circumflex over (b)} for each receivedburst can be accomplished using various numerical search methods such as(for example):$\frac{b_{i}}{a_{i}} = {\frac{b_{i - 1}}{a_{i - 1}} + {f\left\lbrack {Q\left( \frac{b_{({i - 1})}}{a_{({i - 1})}} \right)} \right\rbrack}}$where the function$f\left\lbrack {Q\left( \frac{b_{({i - 1})}}{a_{({i - 1})}} \right)} \right\rbrack$depends on the numerical search method utilized to find the optimal aand b values. For instance, the well known gradient method can be usedwhere f is basically the derivative of Q (calculated as in Equation 24).

It should be understood that variations in the parameters a and bdepends on quantities such as temperature and aging, implying a timeconstant which is much slower than a time constant of the radio channel.Therefore, the algorithm for finding optimal values of â and {circumflexover (b)} need not be executed for every received burst, but just withinsome certain time intervals such as, for instance, every n:th receivedburst, or every k:th second.

How well the AM interfer is eliminated depends on how large |p_(t)|² iscompared to the noise e_(t) and also the relation between a and b. Assuch, for each particular received burst there is a determination as towhether the standard first signal model S_(t) or the second signal model{tilde over (S)}_(t) gives a higher (e.g., highest) signal-to-noiseratio. The processor 426 selects the first signal model S_(t) or thesecond signal model {tilde over (S)}_(t) based on the estimated channelsfilter taps {tilde over (H)} and Ĥ and some quality parameters such asthe residuals obtained in the channel estimators 424 a and 430 b.

Following is an exemplary way of how the processor 426 can decidewhether to use the first signal model S_(t) or the second signal model{tilde over (S)}_(t). First, compare f_(min) (see equation 24) tog_(min) where: $\begin{matrix}{g_{\min} = {\begin{matrix}\min \\{\hat{h}}_{k}\end{matrix}{\sum\limits_{L = 1}^{N}\quad{\left( {S_{t} - {\sum\limits_{k = 1}^{L}\quad{{\hat{h}}_{k}u_{t - k}}}} \right)^{2}.}}}} & (25)\end{matrix}$If f_(min)<αg_(min) (where α is an application specific designparameter) then select the second signal model {tilde over (S)}_(t)otherwise select the first signal model S_(t). Thereafter, the estimatedchannel taps Ĥ or {tilde over (H)} corresponding to the selected signalmodel S_(t) or {tilde over (S)}_(t) are input to the equalizer 428 thatdecodes the received signal.

A typical value of α is 1 which indicates that one selects the signalmodel {tilde over (S)}_(t) or S_(t) having the higher signal-to-noiseratio. However, empirical results indicate a better performance when anα smaller than 1 (e.g., 0.2-0.95) is utilized, in which case the secondsignal model {tilde over (S)}_(t) must have a significantly bettersignal-to-noise ratio than the first signal model S_(t) before it ischosen.

It should also be understood that the direct conversion receiver (anyembodiment) is capable of operating using only the second signal model{tilde over (S)}_(t), instead of having the processor 426 select whichof the signal models S_(t) or {tilde over (S)}_(t) best fits thereceived signal.

Referring to FIG. 5, there is a block diagram illustrating in greaterdetail a second embodiment of an exemplary direct conversion receiver500 in accordance with the present invention. The direct conversionreceiver 500 is similar to the first embodiment except that instead ofusing the received signal to estimate parameters â and {circumflex over(b)}, the direct conversion receiver 500 uses internally generated testsignals c_(t) and d_(t) to estimate parameters â and {circumflex over(b)}.

To avoid repetition, only the components used to internally generate thetest signals c_(t) and d_(t) in the direct conversion receiver 500 aredescribed, because the direct conversion receivers of the first andsecond embodiments otherwise have basically the same architecture andfunctionality.

The direct conversion receiver 500 includes a Digital Signal Processor(DSP) 502 or an Application-Specific Integrated Circuit (ASIC) operableto digitally generate the waveforms of the base band test signals c_(t)and d_(t). The test signals c_(t) and d_(t) include the received(desired) signal and the AM interfering signal, and can be representedas follows:c _(t) =r _(t) cos(Φ_(t))+p _(t) cos(2πΔft+θ _(t))  (26)d _(t) =r _(t) sin(Φ_(t))+p _(t) sin(2πΔft+θ _(t))  (27)where r_(t) cos(Φ_(t)) and r_(t) sin(Φ_(t)) are the I and Q componentsof the desired signal, and p_(t) cos(Δωt+θ_(t)) and p_(t) sin(Δωt+θ_(t))are the I and Q components of the AM interfering signal Δf hertz fromthe desired signal.

The internally generated test signals c_(t) and d_(t) are respectivelyfiltered in low pass filters (LPFS) 504 a and 504 b, and input to mixers506 a and 506 b that convert the base band signals c_(t) and d_(t) up tothe carrier frequency using the oscillating signals LO_(I) and LO_(Q)from the local oscillator 416. The mixers 506 a and 506 b output theirrespective internally generated test signals to an adder 508 whichoutputs an internally generated test signal to the band pass filter 406through a switch 510.

Thereafter, the direct conversion receiver 500 operates to estimate theparameters â and {circumflex over (b)} in the same manner as describedabove with respect to the first embodiment, except that the internallygenerated test signals c_(t) and d_(t) are used instead of the signalreceived at the antenna 402. After estimating the parameters â and{circumflex over (b)}, the switch 510 is positioned to connect theantenna 402 and the band pass filter 406 to enable the further operationof the direct conversion receiver 500 (see discussion with respect toFIG. 4).

An advantage of using the internally generated test signals c_(t) andd_(t) in this self test option is that one can design and control theinterfering signal in such a way that the identification process of theparameters a and b can be easily optimized.

Referring to FIG. 6, there is a block diagram illustrating in greaterdetail a third embodiment of an exemplary direct conversion receiver 600in accordance with the present invention. The direct conversion receiver600 is similar to the first embodiment except that instead of using thereceived signal to estimate parameters â and {circumflex over (b)}, thedirect conversion receiver 600 uses a single internally generated testsignal g_(t) to estimate parameters â and {circumflex over (b)}.

To avoid repetition, only the components used to internally generate thetest signal g_(t) in the direct conversion receiver 600 are described,because the direct conversion receivers of the first and thirdembodiments otherwise have basically the same architecture andfunctionality.

The direct conversion receiver 600 includes a Digital Signal Processor(DSP) 602 or an Application-Specific Integrated Circuit (ASIC) operableto digitally generate the waveform of the base band test signal g_(t).The test signal g_(t) includes the received (desired) signal and the AMinterfering signal, and can be represented as follows:g _(t) =r _(t) cos(Φ_(t))+p _(t) cos(2πΔft+θ _(t))  (28)where r_(t) cos(Φ_(t)) is the I component of the desired signal, andp_(t) cos(Δωt+θ_(t)) is the I component of the AM interfering signal Δfhertz from the desired signal.

Thereafter, the internally generated test signal g_(t) is filtered by alow pass filter (LPF) 604, and input to a mixer 606 that converts thebase band signal g_(t) up to the carrier frequency using the oscillatingsignal LO_(I) from the local oscillator 416. The mixer 606 creates twointerfering signals an equal distance (±Δf) from the desired carriersuch that the use of the desired signal r_(t) cos(Φ_(t)) may not beneeded. Therefore, the internally generated test g_(t) can berepresented as follows:g _(t) =p _(t) cos(2πΔft+θ _(t))  (29)

The mixer 606 outputs the internally generated test signal to the bandpass filter 406 through a switch 610. Thereafter, the direct conversionreceiver 600 operates to estimate the parameters â and {circumflex over(b)} in the same manner as described above with respect to the firstembodiment, except that the internally generated test signal g_(t) isused instead of the signal received at the antenna 402. After estimatingthe parameters â and {circumflex over (b)}, the switch 610 is positionedto connect the antenna 402 and the band pass filter 406 to enable thefurther operation of the direct conversion receiver 500 (see discussionwith respect to FIG. 4).

An advantage of using the internally generated test signal g_(t) in thisself test option is that one can design and control the interferingsignal in such a way that the identification process of the parameters aand b can be easily optimized.

Referring to FIG. 7, there is a block diagram illustrating in greaterdetail a fourth embodiment of an exemplary direct conversion receiver700 in accordance with the present invention. The direct conversionreceiver 700 is similar to the first embodiment except that instead ofselecting the first or second signal model S_(t) or {tilde over (S)}_(t)before the equalizer 428 (see FIG. 4) the selection of the signal modelS_(t) or {tilde over (S)}_(t) is made after first and second equalizers728 a and 728 b.

The direct conversion receiver 700 includes an antenna 702 for receivinga signal from a transmitter 704. The received signal is filtered by aband pass filter (BPF) 706 designed to pass a desired frequency bandsuch as the GSM (Global System for Mobile Communications) frequency bandfrom the received signal. The received signal output from the band passfilter 706 can be represented as: $\begin{matrix}\begin{matrix}w_{t} & = & {y_{t} + \overset{\sim}{p_{t}}}\end{matrix} & (30) \\\begin{matrix}\quad & {\quad =} & {{r_{t}{\cos\left( {{\omega_{0}t} + \varphi_{t}} \right)}} + {p_{t}{\cos\left( {{\omega_{1}t} + \theta_{t}} \right)}}}\end{matrix} & (31)\end{matrix}$where w_(t) is the received signal, y_(t) is the desired signal atcarrier frequency ω₀, {tilde over (p)}_(t) is the near-channel AMinterfering signal at carrier frequency ω₁, and ω₀ and ω₁ are within thepass band of the band pass filter 706.

The filtered signal is amplified in a low noise amplifier (LNA) 708 anddown-converted to a base band Inphase (I) component and a base bandQuadrature (Q) component using mixers 714 a and 714 b, respectively, anda local oscillator (LO) 716. The local oscillator 716 outputs twooscillation signals LO_(I) and LO_(Q) adapted to a carrier frequency ofthe received signal, the two oscillation signals can be represented asfollows:LO _(I)(t)=cos(ω₀ t)  (32)LO _(Q)(t)=sin(ω₀ t)  (33)where LO_(I) and LO_(Q) are the oscillating signals associated with theI and Q components, respectively. The oscillating signals LO_(I) andLO_(Q) and the received signal are multiplied in the mixers 714 a and714 b.

Due to the nonlinearities of the local oscillator 716 and interfererleakage (represented with the scale factors α′, β′), the low passfiltered signal output from the I component mixer 714 a and a first lowpass filter 718 a can be represented as follows: $\begin{matrix}\begin{matrix}{{\overset{\sim}{I}}_{t} = {{LPF}\left\{ \left( {{r_{t}{\cos\left( {{\omega_{0}t} + \varphi_{t}} \right)}} + {p_{t}{\cos\left( {{\omega_{1}t} + \theta_{t}} \right)}} +} \right. \right.}} \\\left. {\left. {\alpha_{I}^{\prime}{\cos\left( {{\omega_{0}t} + \gamma} \right)}} \right)*\left( {{\cos\left( {\omega_{0}t} \right)} + {\beta_{I}^{\prime}p_{t}{\cos\left( {{\omega_{1}t} + \theta_{t} + \delta} \right)}}} \right)} \right\}\end{matrix} & (34) \\{\quad{= {{r_{t}\cos\quad\varphi_{t}} + {\beta_{I}p_{t}^{2}} + \alpha_{I}}}} & (35)\end{matrix}$Likewise, the low pass filtered signal output from the Q component mixer714 b and a first low pass filter (LPF) 718 b can be represented asfollows:{tilde over (Q)} _(t) =r _(t) sin φ_(t)+β_(Q) p _(t) ²+α_(Q)  (36)

Thereafter, the I and Q components are respectively converted intodigital domain by analog-to-digital convertors (A/Ds) 720 a and 720 band respectively filtered by second low pass filters (LPFs) 722 a and722 b. And, after certain normalizations the base band components can berepresented as:Ĩ _(t) =I _(t) +a|p _(t)|² +I _(DC)  (37){tilde over (Q)} _(t) =Q _(t) +b|p _(t)|² +Q _(DC)  (38)where I_(t), Q_(t) are the wanted I and Q components, and I_(DC), Q_(DC)are the DC components on the I and Q components, respectively. |p_(t)|²is the low pass filtered and sampled squared envelope of the interferingAM signal. In case of digital transmission over radio channels withintersymbol interference, such as for instance in GSM or a D-AMPScellular systems, the wanted I and Q components can be written in acomplex notation as follows: I _(t) +jQ _(t) =H ^(T) U _(t) +e _(t)  (39)where H=[h₀, . . . , h_(L)]^(T) is a vector of complex valued channelfilter taps, U_(t)=[u_(t), . . . , u_(t-L)]^(T) is a vector of complextransmitted symbols, and e_(t) is some kind of complex valued noise.Therefore, the complex valued base band signal or a first signal modelS_(t) can be represented as: $\begin{matrix}{S_{t} = {{\overset{\sim}{I}}_{t} + {j\quad{\overset{\sim}{Q}}_{t}}}} & (40) \\{\quad{= \left. {{H^{T}U_{t}} + \left( {a + {j\quad b}} \right)} \middle| p_{t} \middle| {}_{2}{{+ {DC}} + e_{t}} \right.}} & (41) \\{\quad{= {{\sum\limits_{k = o}^{L}\quad{\left\{ {h_{k}^{I} + {j\quad h_{u}^{Q}}} \right\}\left\{ {U_{t}^{I} + {j\quad U_{t}^{Q}}} \right\}}} + e_{t}^{I} + {j\quad e_{t}^{Q}}}}} & (42)\end{matrix}$where j=√{square root over (−1)}.

The Ĩ_(t) and {tilde over (Q)}_(t) components are input to a firstchannel estimator 724 a that correlates, using the first signal modelS_(t), a known training sequence (TS) with the received signal S_(t)containing the same known training sequence to determine asynchronization position and an estimate of the first set of channelfilter taps Ĥ. The use of the first signal model S_(t) in estimating thechannel filter taps Ĥ is well known in the art. The estimated channelfilter taps Ĥ are then input to a first equalizer 728 a that decodes thesignal and outputs the decided signal û_(t) and metrics to a processor726.

Generally, the present invention includes a second channel estimator 724b that utilizes a second signal model {tilde over (S)}_(t) to estimate asecond set of channel filter taps {tilde over (H)}. The second set ofchannel filter taps {tilde over (H)} are input to the second equalizer728 b that decodes the signal and outputs the decided signal û_(t) andmetrics to the processor 726. To obtain the second signal model {tildeover (S)}_(t) represented as the component${{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}},$the second low pass filter 722 a outputs the Ĩ_(t) component to amultiplicator 730 that also receives estimated parameters {circumflexover (b)}/â from channel estimator 724 b. The estimated parameters â and{circumflex over (b)} are determined in the channel estimator 724 b in asimilar manner as described-above with respect to the channel estimator424 b of the first embodiment (see FIG. 4 and related description). Themultiplicator 730 outputs${- \frac{\hat{b}}{\hat{a}}}{\overset{\sim}{I}}_{t}$to a subtractor 732 that also receives {tilde over (Q)}_(t) and, inturn, outputs${\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}$to the second equalizer 728 b and the second channel estimator 724 b.

More specifically, the second channel estimator 724 b correlates, usingthe second signal model {tilde over (S)}_(t), the known trainingsequence (TS) with the${\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}$component to determine a synchronization position and an estimate of thesecond set of channel filter taps {tilde over (H)}. The estimatedchannel filter taps {tilde over (H)} are input to the second equalizer728 b that decodes the signal and outputs the decided signal ũ_(t) andmetrics to the processor 726. Thereafter, the processor 726 selectseither the first or second signal models S_(t) or {tilde over (S)}_(t)based on the metrics and decided signals û_(t) and ũ_(t).

The selection between the signal models {tilde over (S)}_(t) and S_(t)in this embodiment is based on metrics, more particularly in theequalizers 728 a and 728 b the respective channel filter taps Ĥ and{tilde over (H)} are each used to decode the symbols û and ũ. Themetrics for the second signal model {tilde over (S)}_(t) is similar toequation 24, and the metrics for the first signal model S_(t) is similarto equation 25, but instead of minimizing with respect to the radiochannels {tilde over (H)} and Ĥ one minimizes with respect to thesymbols û and ũ. Thus, the same kind of decision process can be used asin the channel estimation case. For example, if metric (of {tilde over(S)}_(t))<α(metric) (of S_(t)) then use the estimated symbols from thesecond signal model {tilde over (S)}_(t) to further process the receivedsignal; otherwise use the estimated symbols from the first signal modelS_(t) to further process the received signal.

It should be understood that the direct conversion receiver 700 can alsobe adapted to use the internally generated test signals c_(t) and d_(t)(see FIG. 5) or the internally generated test signal g_(t) (see FIG. 6)to estimate the parameters â and {circumflex over (b)}.

Referring to FIG. 8, there is a flowchart illustrating the basic stepsof an exemplary method 800 in accordance with the present invention.Beginning at step 802, the first signal model S_(t) is used to estimatethe set of first channel filter taps Ĥ.

At step 804, the second signal model {tilde over (S)}_(t) is used toestimate the set of second channel filter taps {tilde over (H)},including estimating the a and b parameters using an optimizingalgorithm such as described above with respect to the first embodiment.The received signal, the internally generated signals c_(t) and d_(t)(see FIG. 5) or the internally generated signal g_(t) (see FIG. 6) canbe used to estimate the parameters â and {circumflex over (b)}.

At step 806, a selection of the first signal model S_(t) or the secondsignal model {tilde over (S)}_(t) is made depending on which modelbetter enables the elimination of the near-channel interfering signalfrom the received signal (see description associated with the firstembodiment). The selection of the first signal model S_(t) or the secondsignal model {tilde over (S)}_(t) can take place before the equalizationof the received signal (see FIG. 4) or after parallel equalizations ofthe received signal (see FIG. 7).

In the event the selection of the signal models S_(t) or {tilde over(S)}_(t) takes place before the equalization of the received signal, theselection is made using the estimated plurality of first channel filtertaps Ĥ, the estimated plurality of second channel filter taps {tildeover (H)} and at least one quality parameter. Otherwise, in the eventthe selection of the signal models S or {tilde over (S)}_(t) takes placeafter parallel equalizations of the received signal, the selection ismade using metrics and the decided signals û_(t) and ũ_(t).

At step 808, the received signal is decoded and further processed usingthe selected signal model S_(t) or {tilde over (S)}_(t).

From the foregoing, it can be readily appreciated by those skilled inthe art that the present invention provides an apparatus and method thatcompensates for the problematic time-varying DC offset by effectivelyeliminating the AM interferer from the received signal. Also, theapparatus and method disclosed can suppress the AM interferer in a costand current efficient manner as compared to the prior art. It will alsobe apparent to workers in the art that the invention can be readilyimplemented, for example, by suitable modifications in software,hardware or both, in conventional radio receivers such as directconversion receivers.

Although several embodiments of the method and apparatus of the presentinvention have been illustrated in the accompanying Drawings anddescribed in the foregoing Detailed Description, it will be understoodthat the invention is not limited to the embodiments disclosed, but iscapable of numerous rearrangements, modifications and substitutionswithout departing from the spirit of the invention as set forth anddefined by the following claims.

1. An apparatus capable of compensating for a time-varying DC offset bysubstantially removing an amplitude modulated interfering signal from areceived signal, said apparatus comprising: a first channel estimatorfor estimating a plurality of first channel filter taps using thereceived signal and a first signal model; a second channel estimator forestimating a plurality of second channel filter taps using the receivedsignal and a second signal model; and a processor for selecting which ofthe first signal model and the second signal model operates better tosubstantially remove the amplitude modulated interfering signal from thereceived signal.
 2. The apparatus of claim 1, wherein said first signalmodel is represented as:S _(t) =Ĩ _(t) +j{tilde over (Q)} _(t) where Ĩ_(t) is a baseband inphase(I) signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signalof the received signal.
 3. The apparatus of claim 1, wherein said secondsignal model is represented as:${{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}}$where â and {circumflex over (b)} are estimated parameters of theamplitude modulated interfering signal, and Ĩ_(t) is a baseband inphase(I) signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signalof the received signal.
 4. The apparatus of claim 3, wherein saidestimated parameters â and {circumflex over (b)} are determined inaccordance with an optimizing algorithm.
 5. The apparatus of claim 4,wherein said optimizing algorithm is executed during each receivedburst, during a predetermined number of received bursts or during apredetermined number of seconds.
 6. The apparatus of claim 3, whereinsaid estimated parameters â and {circumflex over (b)} are estimatedusing the received signal.
 7. The apparatus of claim 3, wherein saidestimated parameters â and {circumflex over (b)} are estimated using atleast one internally generated test signal.
 8. The apparatus of claim 1,wherein said processor operates to select the first signal model or thesecond signal model depending on which of the signal models has a highersignal-to-noise ratio.
 9. The apparatus of claim 1, wherein saidprocessor operates to select the first signal model or the second signalmodel using the estimated plurality of first channel filter taps, theestimated plurality of second channel filter taps and at least onequality parameter.
 10. The apparatus of claim 1, further comprising anequalizer for processing the received signal using the selected signalmodel.
 11. The apparatus of claim 1, further comprising a firstequalizer for receiving the estimated plurality of first channel filtertaps and a second equalizer for receiving the estimated plurality ofsecond channel filter taps, wherein said processor operates to receiveinformation from said first equalizer and said second equalizer prior toselecting the first signal model or the second signal model.
 12. Theapparatus of claim 1, wherein said apparatus is a mobile phone, a basestation or a direct conversion receiver.
 13. A communications systemcapable of substantially eliminating a near-channel interferer from areceived signal, said communications system comprising: a first channelestimator for estimating a plurality of first channel filter taps usingthe received signal and a first signal model; a second channel estimatorfor estimating a plurality of second channel filter taps using thereceived signal and a second signal model; and a processor for selectingwhich of the first signal model and the second signal model operatesbetter to substantially eliminate the near-channel interferer from thereceived signal.
 14. The communications system of claim 13, wherein saidfirst signal model is represented as:S _(t) =Ĩ _(t) +j{tilde over (Q)} _(t) where Ĩ_(t) is a baseband inphase(I) signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signalof the received signal.
 15. The communications system of claim 13,wherein said second signal model is represented as:${{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}}$where â and {circumflex over (b)} are estimated parameters of thenear-channel interferer, and Ĩ_(t) is a baseband inphase (I) signal and{tilde over (Q)}_(t) is a baseband quadrature (Q) signal of the receivedsignal.
 16. The communications system of claim 15, wherein saidestimated parameters â and {circumflex over (b)} are determined inaccordance with an optimizing algorithm.
 17. The communications systemof claim 16, wherein said optimizing algorithm is executed during eachreceived burst, during a predetermined number of received bursts orduring a predetermined number of seconds.
 18. The communications systemof claim 15, wherein said estimated parameters â and {circumflex over(b)} are estimated using the received signal.
 19. The communicationssystem of claim 15, wherein said estimated parameters â and {circumflexover (b)} are estimated using at least one internally generated testsignal.
 20. The communications system of claim 13, wherein saidprocessor operates to select the first signal model or the second signalmodel depending on which of the signal models has a highersignal-to-noise ratio.
 21. The communications system of claim 13,wherein said processor operates to select the first signal model or thesecond signal model using the estimated plurality of first channelfilter taps, the estimated plurality of second channel filter taps andat least one quality parameter.
 22. The communications system of claim13, further comprising an equalizer for processing the received signalusing the selected signal model.
 23. The communications system of claim13, further comprising a first equalizer for receiving the estimatedplurality of first channel filter taps and a second equalizer forreceiving the estimated plurality of second channel filter taps, whereinsaid processor operates to receive information from said first equalizerand said second equalizer prior to selecting the first signal model orthe second signal model.
 24. A method of reducing an effect of anear-channel interfering signal on a received signal, comprising thesteps of: utilizing a first signal model and the received signal toestimate a plurality of first channel filter taps; utilizing a secondsignal model and the received signal to estimate a plurality of secondchannel filter taps; and selecting which of the first signal model andthe second signal model operates better to substantially remove anear-channel interfering signal from the received signal.
 25. The methodof claim 24, wherein said first signal model is represented as:S _(t) =Ĩ _(t) +j{tilde over (Q)} _(t) where Ĩ_(t) is a baseband inphase(I) signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signalof the received signal.
 26. The method of claim 24, wherein said secondsignal model is represented as:${{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}}$where â and {circumflex over (b)} are estimated parameters of thenear-channel interfering signal, and Ĩ_(t) is a baseband inphase (I)signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signal ofthe received signal.
 27. The method of claim 26, wherein said step ofutilizing a second signal model further includes estimating saidparameters using an optimizing algorithm.
 28. The method of claim 27,wherein said step of estimating said â and {circumflex over (b)}parameters occurs during each received burst, during a predeterminednumber of received bursts or during a predetermined number of seconds.29. The method of claim 26, wherein said estimated parameters â and{circumflex over (b)} are estimated using the received signal or atleast one internally generated test signal.
 30. The method of claim 24,wherein said step of selecting the first signal model or the secondsignal model is determined using the estimated plurality of firstchannel filter taps, the estimated plurality of second channel filtertaps and at least one quality parameter.
 31. The method of claim 24,further comprising the step of decoding the received signal using theselected signal model.
 32. The method of claim 24, further comprisingthe steps of decoding the received signal using the first signal modeland decoding the received signal using the second signal model prior toselecting the first signal model or the second signal model.
 33. Anapparatus capable of substantially removing an amplitude modulatedinterfering signal from a received signal, said apparatus comprising: achannel estimator for estimating a plurality of first channel filtertaps using a signal model represented as:${{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}}$where â and {circumflex over (b)} are estimated parameters of theamplitude modulated interfering signal, and Ĩ_(t) is a baseband inphase(I) signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signalof the received signal; and an equalizer, coupled to said channelestimator, for processing the received signal using the estimatedchannel filter taps.
 34. A method for substantially removing anamplitude modulated interfering signal from a received signal, saidmethod comprising the steps of: estimating a plurality of first channelfilter taps using a signal model represented as:${{\overset{\sim}{S}}_{t}\left( \frac{\hat{b}}{\hat{a}} \right)} = {{\overset{\sim}{Q}}_{t} - {\frac{\hat{b}}{\hat{a}}{\overset{\sim}{I}}_{t}}}$where â and {circumflex over (b)} are estimated parameters of theamplitude modulated interfering signal, and Ĩ_(t) is a baseband inphase(I) signal and {tilde over (Q)}_(t) is a baseband quadrature (Q) signalof the received signal; and processing the received signal using theestimated channel filter taps.